Tech Talks and Tips by K4KYV

Discussion in 'Amplitude Modulation' started by N6YW, Jun 6, 2016.

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  1. K4KYV

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    Possibly a good Class-B driver tube or replacement
    One of the better tubes for class-B audio driver service is the 6B4-G or its equivalent, the 2A3. Unfortunately, audiophools have driven the prices of these tubes sky-high, and once relatively plentiful, they are now practically unobtanium at hamfests. A possible substitute may be the 6CK4, a tube originally designed for service as vertical deflection amplifiers in TVs. The tube has characteristics that would appear to make it nearly ideal for class-B driver service: low amplification factor (6.6), low plate resistance (1200 ohms), high DC cathode current (100 mA, 350 mA peak) and high plate voltage (550 volts maximum DC, up to 2000 volts peak).

    The plate dissipation is slightly lower than that of the 6B4G (12 watts versus 15 watts), and the plate resistance is higher (1200 ohms versus 800 ohms), but two or three pairs of these tubes could be used in a push-pull/parallel circuit, and the higher maximum DC plate voltage rating (550 volts DC versus 300 volts) might allow them to more fully swing the grids of class-B modulator tubes than would 6B4s. The 6CK4 would appear to be superior to a triode-connected 6L6 (plate resistance 1200 ohms versus 1700 ohms for the triode 6L6, 5500 micromho transconductance versus 4700 for the triode 6L6, and even slightly better than the 5250 mircomhos for the 6B4G).

    Availability of these tubes could be a problem, since like sweep tubes, it was manufactured for tube type TV sets, but if you can find them they may go for a better price, as tube vendors apparently think the 6B4G and 2A3 are made of solid gold. I haven't checked out this tube on the tube vendor sites, but they are reported to still be available for around $10-12 apiece. Of course, to use any of these tubes, there is the dreaded problem of the driver transformer, but 6B4Gs and 6L6s were used in several 250-watt class broadcast transmitters so parted-out transformers may be available, or the 6CK4 could be used as a substitute in an existing converted BC transmitter.

    Here are links to characteristics of the 6CK4, 6B4G/2A3 and triode-connected 6L6

    http://tubedata.milbert.com/sheets/093/6/6CK4.pdf

    http://tubedata.milbert.com/sheets/049/2/2A3.pdf

    http://tubedata.milbert.com/sheets/049/6/6L6.pdf
     
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  2. K4KYV

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    Coil winding off-by-one error

    Commonly referred to as the fencepost error and occasionally called a "telegraph pole" or "lamp-post" error. If you run an elevated feed-line, 100 ft. long with supports 10 ft. apart, how many supports do you need? The intuitive answer 10 is wrong. The line has 10 elevated sections, but 11 supports as illustrated below.

    [​IMG]

    This error could just as well be called the coil turns error, since it may lead to confusion when counting the number of turns on a coil. Using the diagram above, you start at the beginning of the coil, count turns along the edge to the other end, and total is 11. But the total number of turns is not 11; it is only ten, since you have to complete the first turn before you tick off number one, which ends at the second strand of wire that you encounter while counting, not the first. To avoid this error, count the first wire at the very beginning of the coil as the zeroth turn.
     
  3. K4KYV

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    Peak, Average and RMS
    Maybe the following explanation will clear up some of the confusion over the terms peak, average and RMS voltage and current, peak versus average power, why RMS voltage and current are not the same as average voltage and current, and why average power is not calculated by multiplying average voltage times average current.

    The root-mean-square (RMS) value of an alternating voltage or current is defined as the equivalent DC voltage or current that would deliver the same amount of energy to a resistor as the AC does over a complete cycle.

    The formulae given in the textbooks for alternating currents assume a sine wave. The sine waveform as seen on an oscilloscope can be thought of as the projection of a circular rotation, viewed edge-on as it moves steadily across the field of vision, to a two-dimensional plane. Mathematically this is called a “rotating vector”. This is where all the trigonometric calculations involving angles, degrees and π (pi) come from.

    Another term for R.M.S. is the quadratic mean. That is, the mean, or average, of a varying quantity taken to the second power. Why the second power (square)? Simple Ohm's law. The power dissipated in a resistor is equal to the square of the voltage across the resistor, divided by the resistance (V²/R), or the square of the current through the resistor multiplied by the resistance (I²R).

    For to-day’s digitally-minded, this might be easier to visualise if we consider a pulsating DC voltage varying between some fixed value and zero, instead of a continually varying sinusoidal AC voltage for which we have to maintain the mental image of the rotating vector simultaneously to pondering all the peaks, averages, squares and square roots.

    Imagine that we have a battery that puts out exactly 2 volts, a 1Ω resistor and a SPST switch, as shown in the attached diagram. Imagine someone with superhuman reflexes opening and closing the switch precisely every millisecond, so that we have a series of square pulses, each exactly one millisecond in length, separated by a space of one millisecond. Or think of the switch as an electronic keyer with the dit paddle held down and the speed turned up to 500 dits per second. See the attached drawing.

    When the switch is closed, the voltage across the resistor is 2 volts. That would be the peak voltage. But since the switch is open 50% of the time, the average voltage across the resistor is only 1 volt. Connect an electromechanical analogue DC voltmeter across the resistor and it will read exactly 1 volt. Likewise, when the switch is closed, the peak current through the resistor is 2 amps, but with the 50% duty cycle, the average current is 1 amp.

    So how hot does the resistor get? Look at the power dissipated in the resistor. While the switch is closed, we have 2 volts across 1 ohm. As stated above, power (P) = V²/R. Therefore we are dissipating 4 watts in the 1Ω resistor while the switch is closed. This is the peak power. But since the switch is open 50% of the time, the average power dissipated in the resistor is 2 watts.

    As we have already noted, the average voltage = 1 volt, and the average current = 1 amp. Multiplying average voltage times average current would give us 1 watt, but we have already shown that the average power is 2 watts. Average voltage × average current yields a physically meaningless figure.

    Now, what is the RMS voltage across the resistor? The equivalent steady DC voltage that would deliver the same quantity of energy per second (average power) to the resistor as does our pulsating DC, can be calculated, since we know the average power dissipated in the resistor. What steady DC voltage would get our resistor just exactly as hot as our pulsating 2 volts? Rearranging the equation P = V²/R gives us V² = PR. We have shown that the average power is 2 watts and our resistor is 1 ohm. So V² = 2 watts × 1Ω, or 2 volts². The RMS voltage, that is, equivalent steady DC voltage that would have the same heating effect on our resistor is the square root of V²: Vrms = √2, or 1.414 volts.

    Thus, we see that our peak voltage is 2 volts, our average voltage is 1 volt and our RMS voltage is 1.414 volts.

    Our peak power is 4 watts, and our average power is 2 watts. Note that the term "RMS power" is not used. Just like multiplying average voltage × average current, the RMS figure for a varying power level can be calculated mathematically, but would yield a meaningless number. There is no such thing as RMS power!

    Where does the term “RMS” (root means square) come from? In the circuit described above we simplified matters by using a 1Ω resistor and a square wave, so that power (P) is simply equal to the voltage squared over a 50% duty cycle. For any waveform, from a simple sine wave or square wave to the most complex waveform imaginable, RMS is the square-Root of the Mean (average value) of the Squares of the varying quantity, as taken incrementally through one complete cycle.


    avepeakrms.png






     
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  4. K4KYV

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    Signal Bandwidth and "Wide" Reports

    It is not unusual to be engaged in an AM QSO and have a SSB station attempt to break in and report that the signals are "wide" or "splattering all over the band". Most of the time, I take those reports with a grain of salt.

    Many operators seem to be unaware that the apparent bandwidth of a signal as it is tuned in on a receiver is the sum of the transmitted bandwidth and the width of the passband selectivity of the receiver. For example, when tuning in a CW signal or unmodulated carrier using a receiver with the selectivity set to 3 kHz in SSB mode, the signal will appear across 3 kHz of dial space on the receiver, even though a clean unmodulated carrier has zero bandwidth. A 3 kHz wide SSB signal tuned in on the same receiver will appear across 6 kHz of dial space. A 7 kHz wide AM signal will appear across 10 kHz of dial space.

    If the signal is high in strength and well above the background noise, it will likely appear even wider, since the selectivity passband characteristic of any receiver is less than a perfect rectangle; even the best bandpass filters have some slope at the skirts. In addition, no transmitter is 100% free of spurious distortion products and no receiver is 100% free of spurious responses. If a signal is coming in at 40 dB over S9, even though the spurious products outside the passband are within FCC specifications at -40 dB, these distortion products will still appear as S9 at the receiver!

    The total bandwidth of a signal is a product of two distinct characteristics: the bandwidth (frequency response) of the signal used to modulate, and spurious distortion products. Both characteristics may cause what is deemed excessive "excessive" bandwidth, but spurious distortion products are far more likely than the audio frequency bandwidth to cause harmful interference to adjacent channel communications, particularly when the source of modulation is the human voice.


     
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  5. K4KYV

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    The Physical Reality of Sidebands
    From the beginnings of radiotelephony it has been debated whether sidebands exist as physical reality or only in the mathematics of modulation theory. In the early 20's, a noted group of British engineers maintained that sidebands existed only in the mathematics, while an equally adamant group of American engineers argued that sidebands do, in fact physically exist.

    Today, the issue seems settled once and for all. We can tune our modern-day highly selective receivers through double-sideband and single-sideband voice signals, and tune in upper or lower sideband, and even adjust the selectivity to the point that we can tune in the carrier minus the sidebands. Nearly everyone accepts the notion that sidebands indeed exist physically... but do they?

    Maybe it's a matter of how we observe the signal, and our result is modified by our measuring techniques. Those who have studied quantum mechanics will recall the Heisenberg Uncertainty Principle, which states that it is impossible to know both the position (physical location) and velocity (speed and direction) of a particle at the same time. This principle is similar but not exactly the same as something called as the "Observer Effect", which states that you cannot observe a system without changing something in the system. In the following thought experiment, let's take this to an analogy with an amplitude modulated radio signal.

    Imagine a cw transmitter equipped with an electronic keyer. Also imagine that there is no shaping circuitry, so that the carrier is instantly switched between full output and zero output. Such a signal can be expected to generate extremely broad key clicks above and below the fundamental frequency because of the sharp corners of the keying waveform. Set the keying speed up to maximum, and send a series of dits. If the keyer is adjusted properly, the dits and spaces will be of equal length, identical to a full carrier AM signal 100 percent modulated by a perfect square wave.

    Suppose the keyer is adjusted to send, say, 20 dits per second when the "dit" paddle is held down. The result is exactly the same thing as an AM signal, modulated 100% with a perfect 20 Hz square wave. Now turn the speed up. If the keyer has the capability, run it up to 100 dits per second. If you tune in the signal using a receiver with very narrow selectivity (100 Hz or less, easily achievable using today's technology), you can actually tune in the carrier, and then as you move the dial slightly you can tune in sideband components 100, 300, 500 Hz, etc. removed from the carrier frequency. A square wave consists of a fundamental frequency plus an infinite series of odd harmonics (third, fifth, seventh, etc.) of diminishing amplitude. Theoretically you would hear carrier components spaced every 200 Hz throughout the spectrum all the way to visible light. In a practical case, due to the finite noise floor, the diminishing amplitude of the sideband components and selectivity of the tuned circuits in the transmitter tank circuit and antenna itself, these sideband components eventually become inaudibly buried in the background noise as the receiver is tuned away from the carrier frequency.

    Suppose we now gradually slow down the keyer. As we change to lower keying speed, it takes more and more selectivity to discriminate between carrier and sideband components, as the modulation frequency becomes lower and the sideband components become spaced more closely together. Let's observe what happens when we slow the dit rate down to 10 dits per second. Now the fundamental modulation frequency is 10 Hz, and we can hear sideband components at 10 Hz, 30 Hz, 50 Hz, 70 Hz removed from the carrier, continuing above and below the carrier frequency at intervals of 20 Hz until we reach a point where the signals disappear into the background noise. In order to distinguish individual sideband components, we need selectivity on the order of 10 Hz, which is possible if we use resonant i.f. selectivity filters with extremely high "Q". This can be accomplished using analogue methods such as crystal filters, regenerative amplifiers or even conventional L-C tuned circuits if we carefully design the components to have high enough Q.

    As we achieve extreme selectivity with these high Q resonant circuits, we observe a sometimes annoying characteristic familiarly known as "ringing." This ringing effect is due to the "flywheel effect" of a tuned circuit, the same "flywheel effect" that allows the abrupt pulses of a class-C tube type final to generate a harmonic-free sine-wave rf carrier waveform. The selective RF tank circuit stores energy which is re-released to fill in missing parts of the sine-wave, thus filtering out the harmonics inherent to operation of this class of amplifier. CW operators are very aware of the ringing effect of very narrow crystal and mechanical filters, which can make the dits and dahs of high speed CW run together, causing the signal to be just as difficult to read with the narrow filter in line, as the same CW signal would be with a wider filter that admits adjacent channel interference. Kind of a damned if you do, damned if you don't scenario.

    Now, let's continue with our thought experiment, taking our example of code speed and selectivity to absurdity. We can slow down our keyer to a microscopic fraction of a Hertz, to the point where each dit is six months long, and the space between dits is also six months long. In effect, we are transmitting an unmodulated CW carrier for six months, then shutting down the transmitter for six months. But still, this is only a matter of degree of sending speed; the signal waveform is still identical to the AM transmitter tone modulated with a perfect square wave, but whose frequency is one cycle per year, or 3.17 × (10 to the -8) Hz. That means that in theory, the steady uninterrupted carrier is still being transmitted, along with a series of sideband components spaced away from the carrier every 6.34 × (10 to the -8) Hz.

    Now, carriers spaced every 6.34 × (10 to the -8) Hz apart are inarguably VERY close together, to the point that building an analogue filter capable of separating them would likely be of complexity beyond the order of a manned expedition to Mars, but still theoretically possible. Let us assume we have developed the means to build such a filter. We would probably resort to superconductivity in the tuned circuits, requiring components cooled to near absolute zero, and thoroughly shield every rf carrying conductor to prevent radiation loss, but here we are talking about something hypothetical, without the practical restraints of cost, construction time and availability of material, so let us just assume we were able to successfully build the required selectivity filter.

    The receiver would indeed be able to discriminate between sidebands and carrier of the one cycle/year or 3.17 × (10 to the minus 8) Hz modulated AM signal, identical to a CW transmitter with carrier on for six months and off for six months. So how can we detect a steady carrier while the transmitter is shut off for six months? The answer lies in our receiver. In order to achieve high enough selectivity to separate carrier and sideband components at such a low modulating frequency and close spacing, the Q of the tuned circuit would have to be so astronomically high that the flywheel effect, or ringing of the filter, would maintain the missing RF carrier during the six-month key-up interval.

    This takes us back to the long-standing debate over the reality of sidebands. If we use a wideband receiver such as a crystal set with little or no front-end selectivity, we can indeed think of the AM signal precisely as a steady carrier that varies in amplitude in step with the modulating frequency. This is always the case if the bandwidth of the signal is negligible compared to the band-pass selectivity of the receiver. Once we achieve narrower selectivity, of the same order as the bandwidth of the signal, which has been the norm for practical receivers dating from the early 1900's up to the present, reception of the signal behaves according to the principle of a steady carrier with distinctly observable upper and lower sidebands. The "holes" in the carrier at the troughs of negative modulation peaks equalling 100%, or 100% modulated with a perfect square wave, are inaudible due to the flywheel effect of the tuned circuits in the i.f. stages of the receiver, even though those same "holes" are still observable on the envelope pattern of an oscilloscope.

    An oscilloscope set up for envelope pattern, with the deflection plates coupled directly to a sample of the transmitter's output, is a non-selective wideband device much like a crystal set, with a selectivity pass-band far exceeding the bandwidth of the signal. It allows us to physically observe the AM signal as a carrier of varying amplitude. A spectrum analyser on the other hand, is an instrument of high selectivity, namely a selective receiver programmed to sweep back and forth across a predetermined band of spectrum while visually displaying the amplitude of the signal falling into its passband at each instant. It clearly displays distinct upper and lower sidebands with a steady carrier in between.

    Furthermore, it has often been observed that the envelope pattern of a signal as displayed from a scope connected to the i.f. output of a distant receiver can be quite different from what is seen on a monitor scope at the transmitter site. This is yet another example of how the pattern is altered (distorted) by the selective components of the receiver, as well as perturbations of the ionosphere.

    In conclusion, there is no absolute yes or no answer to the age-old question whether or not sidebands are physical reality, or exist only in the mathematics of modulation theory. It all depends on how you observe the signal. In a sense, this is analogous to Quantum versus Newtonian mechanics; both visible light and RF radiation have dual properties of wave motion and emission of particles known as photons, depending on how the radiation is observed. So we may conclude that sidebands physically "exist" only if you use an instrument selective enough to observe them. In other words, their existence depends on whether we observe the signal in the time domain or in the frequency domain.

    Remember the Heisenberg Uncertainty Principle and the associated Observer Effect?



     
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  6. K4KYV

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    Efficiency of the AM Linear Amplifier versus Plate Modulated Transmitter


    The efficiency of a linear amplifier is exactly the same whether it is amplifying SSB or DSB AM with carrier. Efficiency is a function of the amplitude of the signal up to the point of saturation of the amplifier tube(s). Theoretically, the efficiency should be more than 60% when the signal level is just under the saturation point.

    At zero signal level, the efficiency is zero, since the tube draws some static plate current but the output is zero. At intermediate points between zero signal and the saturation point, efficiency varies in direct proportion to the amplitude of the signal.

    The reason that the AM linear gets a bad reputation for "inefficiency" is because with the unmodulated carrier, which should reside midway between zero signal and the saturation point, the efficiency is approximately half the maximum peak efficiency. If the peak efficiency is 60% as mentioned above, the unmodulated carrier efficiency is about 30%. This is most obvious with full carrier AM because of the 100% duty cycle of the carrier, whereas with SSB, voice peaks that hit the midway point are of short duration and cause less heating of the final.

    With SSB, the average signal level with the typical human voice (without a lot of processing or overdriving) is about 30% of the amplitude of the maximum peaks. This means that the efficiency of a SSB linear running a clean signal averages something on the order of 18% most of the time even though it may run 60% or more on the maximum peaks. By the same token, the AM linear likewise peaks at close to 60% efficiency on positive modulation peaks while running approximately 30% most of the time. Because of the steady carrier, the wasted power dissipated at the plate(s) of the tube(s) is more obvious when the linear is operating on AM.

    With a sine wave tone modulating the carrier 100%, the tube will run cooler than when there is no modulation of the carrier. A properly operating AM linear draws steady plate current regardless of modulation, therefore the DC input is invariable. With 100% sine wave modulation, we see a 50% increase in total rf output, accounting for the upper and lower sideband energy in addition to the carrier. Since the DC input is the same regardless of modulation, that extra 50% has to come from somewhere, so that means the final runs at higher efficiency to generate that extra power.

    High level plate modulation is usually thought to be much more efficient than linear amplification, for running AM. However, the efficiency advantage of plate modulation is less than might be expected. With plate modulation, extra power is consumed by the modulator tube filaments, modulator tube plate dissipation, and the plate and filament supplies for the substantial sized driver stage required to properly run a class-B modulator. Remember, a high level modulator is nothing more than a linear amplifier operating at audio frequencies, subject to the same inefficiencies as those suffered by the RF linear amplifier. The sideband power from a plate modulated transmitter is generated through a series of two marginally efficient amplifier stages: the high level class B or class AB modulator, followed by the class C RF amplifier. On top of that, the modulation transformer itself is less than 100% efficient, so some of the audio power is wasted before it is delivered to the class C final.

    When considering the total efficiency of the transmitter, i.e., the ratio of energy delivered to the antenna compared to the energy consumed from the a.c. power mains, linear amplification has very close to the same efficiency as plate modulation. The same is true with control grid and screen grid modulation, which operate very similarly to linear amplification.

    When total costs are taken into account, the extra expenditure required for larger tubes with adequate plate dissipation for linear operation is easily offset by the added costs of the modulator tubes, modulation transformer, driver transformer and driver tubes, so the linear amplifier or grid modulated final may actually be cheaper than a plate modulated rig with the same rf output.

    The great advantage of the plate modulated class-C final is ease of tune up and adjustment. Linear amplifiers and grid modulated finals require very critical adjustment of rf grid drive level and antenna coupling (loading adjustment), to assure linear modulation capability up to 100%. With a plate modulated amplifier, if the modulator is adequate to fully modulate the final, a wide range of r.f. grid drive and plate loading will work, with a high degree of tolerance for variations away from nominal adjustments.

    Furthermore, since the class-C final amplifier tube is operating at saturation and small variations in grid drive have little effect on performance, adjustment settings are much more stable than with the AM linear. It's not unusual for the unmodulated carrier level to gradually creep up or down over the course of a QSO when using an AM linear, often due to thermal changes in the exciter circuitry.
     
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  7. K4KYV

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    Ground Radials

    The purpose of the ground plane, radial system or counterpoise is two-fold: (1) It provides a conductive path that diverts returning ground currents away from the lossy earth. In a sense, the ground plane can be thought to "shield" the radiating element of the vertical antenna from the nearby lossy earth. (2) In the case of a Marconi type antenna, it efficiently provides the "missing portion" that would be present in a self-resonant conductor, a half wavelength (or multiple thereof), isolated in space.

    A vertical, inverted-L or tee antenna of an even number of quarter-wave lengths (half-wave, full wave, etc.) functions as a Hertz type antenna similar to an ordinary horizontal dipole rather than as a Marconi, therefore no counterpoise is needed to achieve resonance, but a radial system is still necessary to shield the radiating antenna from the lossy earth. It is a common misconception by hams that no ground system is needed for a half wave vertical. Yes it could be made to resonate, but 50% to 80% of the rf power would be wasted heating the soil in the vicinity of the antenna. Such an antenna used to be called a “worm warmer” by old timers. A radial system is still needed, even with a self-resonant vertical if it is mounted close to the ground. A half-wave vertical may be made to work efficiently if the bottom end is mounted a substantial fraction of a wavelength or more above ground on an insulated pole. The old WWV transmitters on the east coast employed half wave centre-fed vertical dipoles mounted on wooden utility poles.

    The smaller the distance between the ground and the base of the vertical, the more radials are needed to effectively shield the lossy earth from the antenna, since ground currents become significant as the base approaches close proximity to the earth. Ideally, a ground mounted vertical would have at least 60 quarter wave radials, and commercial installations usually have at least 120. OTOH, a VHF ground plane, mounted several wavelengths above ground, may work efficiently with only 3 quarter-wave radials because the vertical radiator is far enough away from the lossy earth that little or no shielding is needed, and the radials serve mainly to complete the "missing half" of the resonant length of the antenna. At intermediate distances between those two extremes, it follows that more radials will be needed as the ground plane approaches the earth, up to the maximum number when the radials are actually in contact with the ground.

    It is not necessary to bury the radials; they may just as well be left lying on the surface for the ground system to function. The primary purpose for burying them is for protection from damage from surface traffic such as wayward feet and lawn mowers. But you don’t want to bury the wires very deeply; an inch or two is enough. Burying the radials more deeply than just slightly below the surface would tend to defeat the whole purpose of the ground plane in the first place, by inserting a layer of lossy earth between the vertical radiator and the ground plane. Avoiding the arduous task of burying thousands of feet of wire, newly-installed radials may be securely fastened to the surface of the ground using clips, stakes or other suitable mechanical means, and they will soon work their way down through the grass and become covered by thatch, and within a year or two will have buried themselves just beneath the sod and may actually be difficult to pull up by hand. Some broadcast stations, with FCC approval, have successfully deployed an elevated radial system, high enough off the ground to allow farm equipment and foot traffic to pass underneath, and experimental data suggests little or no loss of efficiency with far fewer elevated radials than the standard count of 120.

    Beware of those commercially made, ground mounted, electrically short "no-radials-needed" amateur radio verticals; they are nothing more than a bogus rip-off designed to thin the wallets of the blissfully ignorant. The sales pitch would suggest the ability to violate the laws of physics. Better to save your money and construct a real vertical.

    I would recommend studying the seven-part series of articles by Rudy Severns, N6LF, “Experimental Determination of Ground System Performance for HF Verticals” which appeared in January/February 2009 through January/February 2010 issues of QEX. The second instalment discusses a particularly interesting prediction using the modelling program NEC, verified in the experiments, that in ground systems with fewer than about 10 radials buried or lying close to the ground, there may be more ground loss when the radials are longer than about 1/ 8 wavelength, in other words, more copper = more loss! This is counter-intuitive to the logical assumption that making radials too long may be a waste of wire but otherwise causes no harm. The writer suggests that the increase in ground loss with longer radials may be due to the effect of resonances in the radial screen.

    http://rudys.typepad.com/files/qex-ground-systems-part-1.pdf
    http://rudys.typepad.com/files/qex-ground-systems-part-2-1.pdf
    http://rudys.typepad.com/files/qex-ground-systems-part-3.pdf
    http://rudys.typepad.com/files/qex-ground-systems-part-4.pdf
    http://rudys.typepad.com/files/qex-ground-systems-part-5.pdf
    http://rudys.typepad.com/files/qex-ground-systems-part-6.pdf
    http://rudys.typepad.com/files/qex-ground-systems-part-7.pdf
     
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  8. K4KYV

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    Synchronous Detection


    The synchronous detector is nothing more than an ordinary product detector equipped with a phase locked loop to lock the beat frequency oscillator to the original AM carrier.

    While it may be obvious that you would get distortion on a DSB signal from a frequency difference between the original carrier and the inserted carrier (BFO), because heterodyne products of the inserted carrier and the upper sideband would not be the same as heterodyne products of the inserted carrier and lower sideband, it's important to understand that the phase difference would be important as well.

    With any kind of double-sideband signal, coherent detection of the sidebands is required. When a carrier is modulated, the two resulting sidebands have a defined phase relationship with each other. This is explained in most handbooks using simple vector diagrams. Dig up one of the early editions of ARRL's Single Sideband for the Radio Amateur, where they compare the detection process of AM with that of SSB. When demodulating a DSB signal, the resulting audio is a composite of both sidebands, USB and LSB. The audio product from each sideband is a function of the phase relationship between the carrier (or BFO) and that sideband.

    With DSB, the detected (demodulated) audio from the two sidebands must be in proper phase relationship so that they are vectorially additive and therefore reinforce each other. The only way that this can be accomplished with inserted carrier from an external BFO is for the BFO to be precisely on frequency and in phase, or 180 degrees out of phase, with the original carrier.

    One way this is accomplished is to use a phase locked loop to cause the BFO to lock onto the original carrier, which may be reduced up to 20 dB or so, making the DSB reduced-carrier transmitter more power-efficient. It is also possible, using a more complicated PLL configuration known as the Costas Loop, to re-establish the frequency and phase of the injected carrier based on the sidebands only. This makes it possible to fully suppress the AM carrier and transmit only the sidebands. Without some kind of a phase-lock configuration, the only way to satisfactorily demodulate a DSB suppressed carrier signal is to use sufficiently narrow selectivity at the receiver to filter out one sideband and copy only half the DSB signal as SSB. But in this process, 6 dB of signal, and 3 dB of s/n ratio is lost. (We regain 3 dB of s/n ratio with the narrower receiver bandwidth filter.)

    With SSB, the phase of the BFO is of no concern, but for the audio to come out right, the frequency still must be precisely the same as that of the suppressed carrier, although a few Hz of frequency error is usually of little or no consequence with speech transmission (but not the case with music). The phase of the demodulated SSB audio is still a function of the phase of the BFO, but there is no opposite sideband that it has to be in proper phase with, so the phase of the BFO signal is unimportant.

    With normal AM reception using an envelope detector such as the conventional diode circuit, this is less a problem, since the original carrier transmitted as part of the AM signal is used to demodulate the sidebands, and this carrier is inherently on frequency and in proper phase relative to the sidebands - unless the amplitude and/or phase relationship between sidebands and carrier is altered by the ionosphere; then we get the familiar selective fading distortion. That's one of the reasons why a synchronous detector designed to lock the local BFO onto the transmitted AM carrier enhances AM reception even with full carrier AM.

    Interestingly, if you take a DSB AM signal, and rotate the phase of the carrier exactly 90 degrees, relative to the sidebands, you end up with pure phase modulation. The resultant amplitude variations of the carrier, caused by the AM USB and LSB components, are exactly out of phase and cancel each other. So the amplitude modulation nulls out, leaving only the phase modulation of the carrier.

    That's how Armstrong generated his first FM signal in the 1930's. He generated a double-sideband signal at an extremely low carrier frequency, right at the upper limit of the range of human hearing. Then he rotated the carrier 90 degrees, and multiplied the resulting phase modulated signal many times, as necessary to reach the VHF transmit frequency. He then converted the phase modulation to frequency modulation by imposing the appropriate pre-emphasis (frequency response) curve between the audio source and the modulator.

    Getting back to the AM signal, the other reason why the synchronous detector enhances full carrier DSB AM reception is that it allows us the full advantage of the product detector for receiving AM. The vast majority of the "advantage" of SSB over AM is due to the advantage offered by the product detector in the receiver. With properly functioning product detection, the only demodulated audio we hear is heterodyne products between the local BFO and the sidebands, plus noise products within the receiver passband that also heterodyne against the BFO. The very principle of operation of the envelope (diode) detector is intermodulation amongst all signal products that exist within the receiver passband, including the desirable signal as well as all the unwanted interference and noise.

    At the output of the envelope detector appear, in addition to the heterodyne products between the carrier and the two sidebands plus noise as mentioned above, heterodyne products between every element of noise and interference falling within the passband and every element of signal that make up each of the sidebands, plus heterodyne products between each element of noise/interference and every other element of noise/interference falling within the passband. All these additional extraneous intermodulation products resulting from noise and interference products heterodyning with each other and with each element of the desired sidebands, produce an additional jumble of noise that degrades the signal-to-noise ratio of the received signal.

    With the synchronous detector, the noise and interference are still heard, but the product detection process adds transparency. The interference does not intermodulate with the desired signal, allowing it to come through more clearly.
     
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  9. K4KYV

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    Purely Resistive versus Highly Reactive Loads: a mechanical analogy.
    This occurred to me recently, while I was attempting to nail back a piece of wood siding that had come loose on an outbuilding.

    Whenever I attempted to drive a nail through the plywood sheathing directly over a stud, the nail went firmly and soundly into the wood, with little effort. But between the studs, the sheathing is springy and has "give" to it, and most of the energy from the hammer tends to momentarily displace the wood instead of driving the nail. The wood reflects the energy back to the hammer and it bounces. Often one ends up bending the nail and/or splitting the wood and sometimes mashing one's finger instead of sinking the nail into the wood.

    It occurred to me that nailing directly over solid studs is analogous to feeding RF energy into a purely resistive load. The energy exerted by the hammer simply overcomes the resistance of the wood to the nail, and the nail is forced into the wood, with little reaction or bounce. But over the springy part between studs, the sheathing "reacts" to the blows of the hammer, and "reflects" most of the momentum back to the carpenter. With each blow the nail may go slightly deeper into the wood, but most of the energy is reflected back to the sender. If enough energy is reflected back unexpectedly, it can cause injury.

    Hammering over a stud is analogous to what happens when a properly matched feedline is terminated into a purely resistive load: no energy is reflected back; it all goes to heat up the resistance or in the case of an antenna, radiate into space. Hammering between studs is analogous to terminating the feed line into a highly reactive load. Part of the energy is reflected back and only a portion goes into the load.

    Although a reactive load (R + xj) may be normal in certain situations, for example a dipole with tuned feeders or a random length base-fed vertical, undesirable results may occur if the system is not tuned to cancel out the reactance as seen by the transmitter: the final tank may be thrown off resonance causing the tube to overheat, air dielectric capacitors may flash over, and coils and mica capacitors may overheat due to high circulating currents.

    Something that does NOT happen, however; reflected power caused by high SWR on the transmission line does not return the RF energy back to the final and cause the tube plates to overheat. That's an old wive's tale (or maybe I should say an old Hammy Hambone's tale). If the final is dipped to resonance and plate current is close to normal, the final should work just fine even with sky-high SWR, but there may be excessive losses in the transmission line, especially with co-ax. If the final runs abnormally hot, it's likely because the final tank was thrown off resonance by the reactive load, just the same as if someone had manually de-tuned the plate capacitor, running the final off dip.
     
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    Restoring Bakelite knobs and dials.

    The lettering or numbering on vintage Bakelite knobs and dials is often deteriorated and in need of restoring. The marks were engraved into the moulding from which the knobs were manufactured, and then filled in with some kind of paint (usually white, but occasionally gold or ivory) after the finished product came out of the mould. The paint tends to eventually dry up and flake out of the engraving, or else becomes dirty and turns dark, or both. Restoration is simple: clean the dirt and crud off the knob, then thoroughly remove and replace the old paint. This must be done with extreme care, however, to avoid permanently damaging the knob or dial

    NEVER clean out the old paint with a sharp metallic object; this will inevitably scratch and visibly damage the moulded impressions. Sharpen a wooden or plastic toothpick, or perhaps a discarded plastic alignment tool. It may help to soften the old markings with paint stripper before attempting to remove them with the tool. From my experience, paint stripper does not damage Bakelite, but it wouldn't be a bad idea to test your stripper on a tiny spot on the back side of the dial first, just in case; manufacturers reformulate chemical products all the time without warning. NEVER NEVER EVER clean those dials with a cleanser like Fantastik or 409! It will wash away the microscopically thin glossy skin from the surface of the bakelite and expose the filler underneath, which is usually made of finely-ground sawdust. Once the filler is exposed, the sheen is lost for ever. Nothing will ever restore it back to original. It is impossible to polish sawdust.

    To restore the markings after stripping down to bare bakelite, I recommend a lacquer stick. These were available in white or gold from Antique Electronic Supply in Tempe, AZ just for this purpose, but I have not checked recently to see if they still have them. If not, do a web search; they should be available elsewhere. Rub the stick over the cleaned-out markings just as you would use a wax crayon, then use a soft cloth to carefully wipe away the paint from the surface of the dial, leaving it embedded in the grooved impressions that form the markings. It may take several attempts to get this right, to completely remove all the paint from the surface while leaving the impressions thoroughly filled in. It may help to dampen (but not saturate) the cleaning rag with mineral spirits. If you can't find a lacquer stick, I have successfully used white oil-based paint and an artist's brush.

    Leave it to dry overnight, and the knob or dial should turn out looking as good as or better than it did originally.

     
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